Two principal goals normally involved in the design of a radio frequency (RF) transmitter are linearity and efficiency. These goals are substantially impacted by the design and operation of the RF transmitter's power amplifier (RF PA). RF PA linearity refers to how capable the RF PA is at producing a linear reproduction of the RF signal applied to its input. RF PA efficiency refers to how energy efficient the RF PA is at converting power supplied to it from its power supply into RF power. Unfortunately, due to physical limitations of the power transistors that make up RF PAs and how the power transistors operate, designing an RF PA that is both highly linear and highly efficient is very difficult.
A third goal often involved in the design of an RF PA is to make the RF PA so that it is capable of operating over a wide range of frequencies and has a wide video bandwidth capability. Designing an RF PA that is capable of operating with high linearity and high efficiency at a specific frequency or over a narrow range of frequencies is difficult by itself. Designing an RF PA that is capable of operating with high linearity and high efficiency over a wide range of operating frequencies and which also has a wide video bandwidth capability is even more, in fact is considerably more, difficult.
To highlight some of the difficulties and challenges involved in designing an RF PA having high linearity, high efficiency, and wide operational and video bandwidth capabilities, reference is first made to FIG. 1, which is a schematic drawing of a conventional RF PA 100 configured for Class-A operation. The Class-A RF PA 100 comprises a power transistor 102 having a collector (or drain, if the power transistor is a field-effect transistor, instead of bipolar junction transistor) configured to receive a power supply voltage VCC. An inductor 104 in the collector path serves as an RF choke that prevents RF energy produced by the RF PA 100 from entering and disrupting the collector power supply. The base of the power transistor 102 is biased by a DC bias voltage VBB, which centers the operating point 202 of the Class-A RF PA 100 near the center of the active region of the I-V characteristic curves of the power transistor 102 (see FIG. 2). The DC bias voltage VBB and collector supply voltage VCC are set, and the peak-to-peak variation of the magnitude of the input RF voltage vin is controlled, so that the RF PA 100 operates about the operating point 202, along a load line 204, exclusively in the active region of the characteristic curves, and preferably never in the saturation and cutoff regions. In this manner, the RF PA 100 is able to produce an RF output voltage vout that is a linear reproduction of the applied input RF voltage vin.
The Class-A RF PA 100 is highly linear. However, unfortunately, it is also highly inefficient. Achieving high linearity (Class-A operation) requires that the RF PA's operating point 202 be positioned near the center of the active region of its power transistor's I-V characteristic curves. Unfortunately, this requirement undesirably results in the power transistor 102 dissipating large amounts of quiescent power. This problem is highlighted in FIG. 3, where it can be seen that the power transistor 102 in a Class-A operation conducts at all times, with a large DC quiescent current ICQ flowing through it at all times, regardless of what the magnitude of the AC component of the collector current iC is. High linearity is therefore obtained at the expense of efficiency. In fact, it can be shown that the maximum realizable conversion efficiency of the conventional Class-A RF PA 100 is only 50%, meaning that at least half the DC power supplied to it from its power supply is dissipated by the power transistor 102 in the form of heat. This low conversion efficiency is highly undesirable, particularly in circumstances where the power supply is a battery, such as in a wireless handset, for example, since the low conversion efficiency substantially limits the charge life of the battery.
One way to avoid the low conversion efficiency problem that plagues the Class-A RF PA 100 is to employ a Class-B RF PA, instead. In a Class-B RF PA the RF PA is biased so that the DC operating point is positioned at the boundary of the active and cutoff regions of its power transistor, as illustrated in FIG. 4, rather than in the center of the active region of the I-V characteristic curves, as in Class-A operation. The benefit of this approach is that the collector current iC is zero when the input RF voltage vin is zero. In other words, the quiescent current in for Class-B operation is ICQ≈0.
One consequence of biasing the RF PA at the boundary of the active and cutoff regions, however, is that the RF PA then only becomes capable of amplifying the positive portions of the RF input. In other words, as illustrated in FIG. 5, the conduction angle of the Class-B RF PA is reduced from a conduction angle of 2π (as in Class-A operation (see FIG. 3)) to π (see FIG. 5). So that the full collector current waveform can be produced at the output, the Class-B RF PA can be constructed using two power transistors—one for amplifying the positive portion of the RF input and the other for amplifying the negative portion. This so-called “push-pull” operation is illustrated in FIG. 6. A typical push-pull RF PA 600 comprises a first power transistor 602 (e.g., first BJT), a second power transistor 604 (e.g., a second BJT), a center-tapped input transformer 606, and a center-tapped output transformer 608. The first and second power transistors 602 and 604 have identical operating characteristics and, as illustrated in the I-V characteristic curves in FIG. 7, are biased by a DC bias voltage VBB at the boundary of their active and cutoff regions for Class-B operation. The bases of the first and second power transistors 602 and 604 are both configured to receive an input RF voltage vin from the secondary winding of the center-tapped input transformer 606, but 180° out of phase. When the RF input to the Class-B RF PA 600 is a sinusoid, the two collector currents iC1 and iC2 produced in the first and second power transistors 602 and 604 are half-wave rectified amplified versions of the sinusoidal input current iin. The collector currents iC1 and iC2 flow through the primary winding of the output transformer 608, where they add together. The summed collector current flowing through the primary winding results in a load current iL being induced in the secondary winding of the output transformer 608. As illustrated in FIG. 8, the load current iL is a full-wave rectified sinusoid. Since the load current iL is a full-wave rectified sinusoid so too then is the RF output voltage vout.
Because ICQ1=ICQ2≈0 in the Class-B RF PA 600, the power transistors 602 and 604 dissipate very little power in the quiescent condition. The Class-B RF PA 600 therefore has a significantly higher conversion efficiency compared to the Class-A RF PA 100. In fact, ignoring power dissipated by its biasing circuits and other non-ideal losses, it can be shown that the Class-B RF PA 600 obtains a maximum conversion efficiency of η=Pout(RF)/PDC=π/4=78.5%.
Although the Class-B RF PA 600 has a maximum conversion efficiency of η=π/4, this maximum conversion efficiency can only be approached in situations where the peak amplitude of the RF input voltage vin does not vary over time, i.e., in situations where the RF input vin voltage has a “constant envelope.” Many modern wireless communications standards such as, for example, Wideband Code Division Multiple Access (WCDMA), Worldwide Interoperability for Microwave Access (WiMAX), Long Term Evolution (LTE) employ nonconstant envelope modulation schemes in which both the amplitude and angle (i.e., phase and/or frequency) of the RF output are varied to increase the amount of data that can be transmitted in a given portion of the RF spectrum. As illustrated in FIG. 9, the signal envelope 902 of an RF output signal RFOUT that has been modulated according to a nonconstant envelope modulation scheme has a nonconstant envelope, with a peak-to-average power ratio (PAPR) determined by the particular modulation scheme being employed. So that the RF PA does not clip the signal peaks of these nonconstant-envelope signals, the RF output power of the RF PA must be backed off by at least the amount of PAPR. Unfortunately, this back-off requirement can substantially limit the realizable conversion efficiency of the Class-B RF PA 600, especially in circumstances where the PAPR is high. In fact, for a PAPR=6 dB, the 6 dB back-off requirement results in the Class-B RF PA 600 only achieving a maximum possible efficiency of less than 40%.
Various prior art approaches have been proposed to increase the low conversion efficiency of the Class-B RF PA 600 when operating at backed-off power levels. One approach is the Doherty RF PA. FIG. 10 is a drawing showing the salient elements of the conventional Doherty RF PA 1000. The Doherty RF PA 1000 comprises a power divider 1002; a carrier (i.e., main) RF PA 1004; a peaking (i.e., auxiliary) RF PA 1006; a first and second quarter-wave impedance inverters 1008 and 1010; and a quarter-wave impedance transformer 1012. The carrier RF PA 1004 is biased for Class-B operation and the peaking RF PA 1006 is biased for Class-C operation. (Note that in Class-C operation the RF PA 1006 is biased fully in the cutoff region, rather than at the boundary of the active and cutoff regions as in Class-B operation, i.e., so that the conduction angle is less than n.) When the Doherty RF PA 1000 is operating at low RF output power levels only the carrier RF PA 1004 is active. The peaking RF PA 1006 remains cut-off and appears as an open circuit to the carrier RF PA 1004. With the peaking RF PA 1006 off, the quarter-wave impedance transformer 1012 and second quarter-wave impedance inverter 1010 together serve to increase the load impedance seen by the carrier RF PA 1004 to R1=2RL=100Ω. The increased load impedance allows the carrier RF PA 1004 to reach full voltage swing at half power. The Doherty RF PA 100 therefore operates twice as efficiently at low RF output power levels than does the conventional Class-B RF PA and at a back-off power level of −6 dB achieves a maximum possible conversion efficiency of η=π/4, as shown in FIG. 11. As the RF input drive to the Doherty RF PA 1000 increases to a value that causes the RF output power to increase above the −6 dB back-off point, the peaking RF PA 1006 begins to turn on and the carrier RF PA 1004 saturates. (Note that the first quarter-wave impedance inverter 1008 is used to compensate for the 90° phase shift of the carrier RF PA 1004 output caused by the second quarter-wave impedance inverter 1010.) As the RF input drive continues to rise above the −6 dB back-off point, the peaking RF PA 1006 injects more and more current into the load RL. Since the carrier RF PA 1004 is saturated above the −6 dB back-off point, if the second quarter-wave impedance inverter 1010 was not to be present the increasing current being injected by the peaking RF PA 1008 would result in the carrier RF PA 1004 seeing an increasing effective load impedance. However, because the second quarter-wave impedance inverter 1010 is present it operates to actually lower the effective load impedance seen by the carrier RF PA 1004, lowering it from the R1=2RL=100Ω maximum value during times when the peaking RF PA 1004 was off to lower and lower values. The lowering effective load impedance seen by the carrier RF PA 1004 allows the carrier RF PA 1004 to increase the amount of power it delivers to the load RL, despite being saturated. The second quarter-wave impedance inverter 1010 also serves to transform the output impedance of the carrier RF PA 1004 to a higher value at its output end, thus allowing the peaking RF PA 1006 to efficiently pump power into the load RL as the input drive continues to rise. At PEP the carrier and peaking RF PAs 1004 and 1006 both see a 50Ω load, each contributes half the overall RF output power, and the overall conversion efficiency of the Doherty RF PA 100 is the same as the non-backed-off conventional Class-B RF PA (i.e., η=π/4). As illustrated in FIG. 11, the load modulation principle exploited by the Doherty RF PA 1000 allows the Doherty RF PA 1000 to achieve a maximum conversion efficiency of η=π/4 at both the −6 dB back-off point and the PEP output point while maintaining a relatively high conversion efficiency in between.
Although the Doherty RF PA 1000 is able to increase conversion efficiencies at backed-off power levels over that which can be realized in the conventional Class-B RF PA 600, it has a number of significant limitations. First, the impedance inverters 1008 and 1010 and impedance transformer 1012 are frequency dependent and consequently limit the possible tuning bandwidth of the Doherty RF PA 1000. These frequency dependent constraints are a significant problem since many modern and evolving communications standards require an RF PA with a tuning bandwidth up to, and in some cases exceeding, 100 MHz. The conventional Doherty RF PA 1000 does not provide this tuning bandwidth capability. Second, although the Doherty RF PA 1000 can achieve higher conversion efficiencies in back-off conditions, to achieve high linearity it requires resonator circuits to filter out unwanted harmonics generated by the nonlinear operation of the Class-B and Class-C carrier and peaking RF PAs 1004 and 1006, and carefully implemented predistortion processes in order to satisfy strict spectral mask requirements. Finally, the Doherty RF PA 1000 is very sensitive to changing load or changing antenna impedances. This sensitivity is a severe problem when applied in mobile handsets, where the effective antenna impedance is influenced by the changing operating environment. Additionally, this load sensitivity of the Doherty RF PA 1000 is problematic in applications intended to support carrier aggregation techniques, such as in the LTE Advanced (LTE-A) mobile communications standard. In such applications some kind of antenna impedance control to handle the total composite signal over the band can have a dramatic impact on the handling of the individual signals yielding undesirable distortion.
Another type of RF PA that can be used to achieve higher conversion efficiencies at backed-off power levels when high PAPR signals are involved is the Class-G RF PA, a simplified drawing of which is provided in FIG. 12. The Class-G RF PA 1200 comprises a first power transistor (e.g., an n-p-n BJT) 1202 and a second complementary power transistor 1204 (e.g., a p-n-p BJT) connected in a complementary push-pull configuration. The DC bias generators 1206 and 1208 serve to bias the first and second power transistors 1202 and 1204, typically for either Class-B or Class-AB operation. The Class-G RF PA 1200 uses a technique known as “rail switching.” As illustrated in FIG. 13, at lower RF output power levels the collectors of the first and second power transistors 1202 and 1204 are configured to receive power supply voltages VCC1 and −VCC1 but at higher RF output power levels are switched to receive power supply voltages VCC2 and −VCC2 of higher magnitudes. Although this rail switching approach can help to improve conversion efficiencies at backed-off power levels, the hard switching of the power supplies yields spurious emissions that are very difficult to correct for.
Yet another approach that can be used to improve conversion efficiency when high PAPR signals are involved is a technique known as envelope tracking (ET). As illustrated in FIG. 14, an ET RF PA 1400 employs a dynamic power supply (DPS) 1402 to supply power to an RF PA 1404. The DPS 1402 produces a DPS voltage VDD(t) that tracks the signal envelope 1406 of the incoming RF signal RFIN. By powering the RF PA 1404 using the DPS voltage VDD(t), rather than using a constant DC supply voltage VDD(DC), the ET RF PA 1400 is theoretically capable of operating at maximum efficiency over all RF output power levels. Although the ET RF PA 1400 is highly efficient, one significant limitation associated with its use relates to the DPS 1402. In situations where the video bandwidth of the signal envelope 1406 is high, the DPS 1402 has difficulty tracking the signal envelope 1406. The reason for the difficulty is that the power transistors that make up the DPS 1402 have practical constraints that limit their ability to react quickly to rapidly changing signal envelopes, especially in circumstances where the PAPR of the signal envelope is high and the magnitude of the signal envelope approaches zero. When the envelope signal bandwidth exceeds the ability of the power transistors to react, significant amplitude distortion results. This practical constraint limits use of the ET RF PA 1400 to applications in which the signal envelope bandwidth is less than about 20-40 MHz.
Considering the drawbacks and limitations of the prior art RF PA approaches summarized above, it would be desirable to have an RF PA apparatus that operates with high linearity and high-efficiency over a wide range of frequencies, has a wide video bandwidth capability, and is insensitive to changing antenna impedances.